Technical Indicators of Devices for Signal Reception and Processing

Abstract

Modern digital telecommunication systems have a complex architecture that includes a fixed network, ground equipment, and a mobile terminal that are part of the radio network structure. The interaction of systems and network equipment, implemented using different principles and technologies and meeting different standards, is provided at the interface level described in the reference model for the interaction of open systems OSI (Open System Interconnect). The radio interfaces of the physical channel are based on the technical indicators of radio receiving devices, including terminal equipment: the type of radio signal modulation, the method of access to the channel, the rate of transmitted signals, the S/N ratio at different points of the radio frequency front end, etc.

These characteristics form the requirements for specific devices that implement the interface and methods (algorithmic and circuitry) providing the specified properties.

The technical specifications for the terminal devices instruct the designers of the RF front-end architecture to meet the requirements for the amount of intermodulation distortion, signal blocking, and cross modulation. The reasons for the occurrence of distortions and ways to reduce them to acceptable values are considered. Simulation in the MicroCap environment of a resistor cascade on BT and calculation of its noise properties are performed.

Appendix to Chapter 5

Application 5.1

Calculation of admissible and real noise factor.

There is the principal difference between receivers with the operating frequencies below 50 MHz and above 50 MHz.

As we see from [117] and from Fig. 5.19, in the first frequency region, the external and atmospheric interferences having the significant intensity act in the receiver input.

Fig. 5.19

The relative level of interferences at the antenna output from various sources

Therefore, in the frequency region below 50 MHz, receivers are under the impact of enough powerful input signals (Fig. 5.20), i.e., they must have the high noise immunity at the relatively low sensitivity.

Fig. 5.20

Dependence of useful power at input receiver on the frequency

In professional receivers of the range of moderate high frequencies, which have a higher sensitivity, SNR value can be chosen significantly lower than for broadcasting receiver and be::]

$$ SN{R}_{in}=\frac{U_{s. in}}{U_{n. in}}=3\div 6\;\mathrm{dB}, $$

where U_{s. in}, U_{n. in} are effective voltage values, relatively, of the signal and noise in the receiver input, owing to this, the sensitivity can be increased.

Evidently the spectral power of external noises (in relative units) acting in the input restricts the minimal power value of the receiving signal. The minimal value of the noise coefficient should correspond to conditions defined by the function “C,” and for systems of personal radio communication, by the function “G” (Fig. 5.19). For determination of the noise temperature in the operating frequency region, which is not presented in Fig. 5.19, we can apply the linear interpolation of the appropriate dependence.

With increasing frequency, the spectral density of external noise decreases (Fig. 5.19), which will reduce the level of signal arriving at the input of the receiver. However, to provide the required SNR at the input of the receiver, it is necessary to reduce its own noise figure (increase the receiver sensitivity), for example, using the spread spectum technology of the received signal. For the case when the noise factor defined by external sources (NF_ext) is equal to the noise factor (real) recalculated to its input (N_real = N_RFfe), NF_ext = NF_RFfe, the noise level in the receiver output becomes higher by 3 dB than at the absence of external noises. In the high-frequency region (up to 30 MHz), it is unreasonable to fight for the reduction of the own noise of the radio path NF_RFfe in order to increase the sesitivity, since the noise generated by external sources and acting at the input of receiver (NF_ext) turns out to be significantly higher then the owen noise at a given SNR. Redicing the level of external noise (and reception of weak signals) can be acheved by using antennas with a narrow radiation pattern and using tunable filters in the preselector, narrowing the received frequency band. In the UHF and microwave region, we must take into account the contribution of Earth’s thermal noises corresponding to temperature Т_n ≈ 290 К^о (NF_ext ≈ 3 dB; Fig. 5.19). For systems of personal communication applying the pencil-beam antennas, the noise factor NF_RFfe should correspond to the level of galactic noises (G; Fig. 5.19), and in practice, it is necessary to achieve the possibly lower values. On frequencies above 200 MHz, the noise temperature Т_n ≈ 50 К^о (NF_ext ≈ 0,7 dB), which allows increasing the receiver sensitivity.

For radio broadcasting receivers operating in the region of moderate frequencies, the main requirement is the high fidelity of reproduction of receiving signals. The fulfillment of this requirement can be achieved at the condition that the minimal signal exceeds the interference level acting in the input by 20–30 dB. Practical estimations of atmosphere and industrial interference level show that for receivers of such a type, the sensitivity should be within the limits: Е_Аmin = 50 ÷ 200 μV, where Е_А is the signal EMF in the receiving antenna.

Achievement of the required single-signal sensitivity at a given SNR is defined, first of all, by the interference power acting in the receiver input and having the various nature and spectral properties as well as by the RF front-end noise power reduced to the input. At development of the receiver with the operation range below 50 MHz, its sensitivity is mainly defined by external interferences (Fig. 5.19), which decreases requirements to noise indices of the receiver itself. With the frequency growth, the level of industrial and natural interferences decreases, and implementation of receivers with increased sensitivity is mainly defined by noise properties of input stages of the radio front end.

When designing the RF front-end, the receiver noise figure (NF_REfe), recalculated to its input (real noise figure), should be no more then the admissible noise figure generated by external sources, including the thermal noise of antenna NF_adm, calculated at its input, usually for conditions corresponding to a quiet area (C, Fig. 5.19). If the condition is met that the permissiblle noise figure is equal to the real NF_adm = NF_RFfe, the noise level at the output of the RF interface will be twice (3 dB) higher then in the absence of external noise. From this, the noise factor of the front end must be selected as: N_RFfe = N_adm – 3, (dB). If we take into account there are losses in the antenna feeder and thermal interference of the Earth is added to the signal receved near the Earth’s surface, then at T = 290 K the permissible value of the noise factor will be NF_adm ≈ 3 dB to fulfillment the condition.

$$ N{F}_{adm}\ge N{F}_{RFfe} $$

(П.5.1)

it is enough to provide the value N_RFfe ≈ 3 dB.

The level of the own noise power (Fig. 5.18) coincides with the power level of intermodulation distortions arising in the front end for useful signal absence in its input due to the impact of out-of-side interferences. The level of the own noise power (Fig. 5.18) coincide with the power level of intermodulation distortions arising in the RF front-end for useful signal absence in its input due to the impact of out-of-side interferences and can be up to ≈ 3 dB. This leads to a decrease in the threshold sensitivity by 3 dB. This factor determines the requirements for the real noise fugure of RF front-end NF_RFfe, which should be close to the value of admissible noise factor NF_adm. Achieving the adequacy of these values (NF_adm ≈ NF_RFfe) is ensured by installing an attenuator directly after the antenna connector, which increases the output at poiint of single-dB compression P1 and the value of the intersection points IP2 and IP3, by reducing the power limit of intermodulation distortion. At the same time, it saves practically unvaried of the dynamic range width on blocking and on intermodulation distortions.

Modern radio communication systems form high requirements to Se_im.ch, which do not permit to realize them using simple preselector constructions. It is clear that it is technologically very difficult to apply tunable filters in the preselector, since even when the AT operates in the standard of a single system, for example, GSM, it is nessesary to provide reception/transmission in the frequency band allocated to the system (about 25 MHz and more) when tuning to subscriber frequency with a relative error of 10 … 10. In addition, the user for work in the VHF, UHH, and SHF band must be provided with the possbility of tuningless operation of the AT recever, which is achived by using quartz-stabilized reference oscillators. Tuning of the AT in frequency when operating in one or several systems is ensured by using frequensy synthesizers that form a frequency grid. Clarification of AT tuning to the received frequency is achived using voltage-controlled generators (VCOs). Such a two-stage system for achieving high accuracy (with a relative error of 10 … 10) of AT tuining to the received signal is used in all data trasmission systems. In systems of the wideband access, the common frequency band is presented to each user; therefore, at saving of high requirements to Se_im.ch, we can apply the pass-band filter usually realized on SAW systems and having the squareness coefficient close to 1 (k_sq,0,001 = 1.2,…,2). Increase of the intermediate frequency simplifies the preselector construction; however, without transfer to the lower intermediate frequency using the second frequency conversion, it is rather difficult to satisfy the requirements to Se_adj.chк and required sensitivity.

Application 5.2

Calculate the frequency of the intermodulation channel of reception of the base station of GSM standard.

The operating frequency band of the base station of the GSM900 standard in the reception mode is 890–915 MHz (Table 1.2). It is assumed that two interfering influences from MS of other operators are simultaneously received at the input of the BS receiver. In this frequency band, we must receive the signal from any mobile station, which supports the active mode with the base station (Fig. 5.21).

Fig. 5.21

Distribution of frequency bands between MS and BS of the GSM system in frequency duplexing mode

The received MS frequency band in FDD mode is 935–960 MHz. In it, two subscribers are allocated a frequency band of 200 kHz for each connection. In this case, the frequency band of the working channel is provided randomly from the number of free channels.

The impact of a certain pair of frequencies emitted by mobile stations of other operators of the same or other systems can form frequency components on the AE of the preselector and frequency convertor the fall into the operating frequency band of the mobile station. The danger of such distortions, called intermodulation distortions, lies in their suddenness and values of affecting frquencies. The following intermodulation distortions of the third order

$$ {f}_{IMD3}=2{f}_1\pm {f}_2, $$

(App. 5.2)

$$ {f}_{IMD3}=2{f}_2\pm {f}_1 $$

(App. 5.3)

can be related to these distortions, where f₁ and f₂ are carrying frequencies in the frequency band of the mobile station transmission (f₁ < f₂). A choice of some values of f₁ and f₂ and their second harmonics from the band of transmitted frequencies can lead to the appearance of f_IMD3 in the band of receiving frequencies of the base station. To reduce the danger of intermodulation distortions’ appearance between allocated bands of reception/transmission, the band of the duplex diversity is allocated, which is 20 MHz for GSM900, in which the operating frequency bands of other systems may be located.

Intermodulation distortion of the third order can occur when conditions (App5.2) or (App. 5.3) are fulfilled. Evidently, the sum sign “+” cannot form components IMD3 in the reception band of the mobile station. The upper limit of the difference combinational component IMD3 from (App. 5.2) cannot exceed limit of the band of received frequencies of the MS, which makes it possible to find the value of the lower frequency f_2IMD3 at which this occurs:

• 2·f₁ – f₂< 960 MHz; from which we obtain: f_2IMD3 > 820 MHz.

From the condition that

• 2·f₁ – f₂ > 935 MHz, it follows: f_2IMD3 < 845 MHz, i.e., 820 < f_2IMD3 < 845 MHz (the region of combinational components). On the frequencies axis, we put the area corresponding to the band of the received of mobile station which is the source of IMD3 (Fig 5.22): (a) frequency band received by MS and formed combination components (from the condition App. 5.2), (b) frequency bands (from the condition App. 5.3)

Fig. 5.22

(a) Frequency bands transmitted by MS (received by BS) and formed combination components (from the condition App. 5.2), (b) frequency bands transmitted by MS (received by BS) and formed combination components (from the condition App. 5.3)

As follows from Fig. 5.22a, the region of components, wich can forme IMD3, is situated outside of the receicing frequency band of mibile stsation, therefore, it is not dandgerous.

Using (App. 5.3), we calculate the frequency band in which there is a danger of IMD3 occurence (Fig. 5.22b):

• 2·f₂ – f₁ > 935 MHz, from which f_{1 IMD3} < 985 MHz;

• 2·f₂ – f₁ < 960 MHz, from which we obtain: f₁ > 960 MHz. In this case, the double-side inequality defined the frequency band of combinational components: 960 < f₁ < 985, MHz, also do not intercepts with the region of operating frequencies (935 … 960 MHz) of mobile station receiver. The reason for this phenomenon, as shown above, is not the influence of high-power out-of-band transmitters, but the interaction of the spectral components of the MS of its own network at the input of the BS receiver, which is a feature of the mobile system. •f - f > 935 MHz, from which f < 985 MHz; 2•f -f < 960 MHz, from which we obtain: f > 960 MHz. In this case, the double-side inequality defined the frequency band of combinational components: 960 < f

Application 5.3

Calculate the value of dynamic range in third harmonic DB3.

Initial data:

• 1) The noise factor of the receiver: NF = 5 dB

• 2) The noise pass-band of the radio front end: B_n = 5 MHz

• 3) The intercept point for intermodulation for the third order on output: IP3 = 50 dBm

Using (5.11) for the noise factor

$$ N=\frac{P_{s, in}/{P}_{n, in}}{P_{s, out}/{P}_{n, out}}, $$

and converting (5.14), we obtain the expression for the noise power produced by an antenna of the mobile station in the receiver input:

$$ {P}_{n, in}=\frac{P_{s, in}}{N\left({P}_{s, out}/{P}_{n, out}\right)}=\frac{S_n\cdotp N\cdotp \left({P}_{s, out}/{P}_{n, out}\right)\cdotp {B}_n}{R_A\left({P}_{s, out}/{P}_{n, out}\right)}=\frac{S_n\cdotp N\cdotp {B}_n}{R_A}, $$

where S_n = 4kT₀R_in is the spectral density of the noise average power.

Assuming that an antenna is matched with the receiver input (R_A = R_in), the thermal noise power is determined as:

$$ \frac{S_n}{R_A}=\frac{4k{T}_0{R}_{in}}{4{R}_A}=k{T}_0. $$

For the room temperature (Т = Т₀ = 290 К), we have:

$$ \frac{S_n}{R_A}=10\lg \left(\frac{k{T}_0}{10^{-3}}\right)=-174\;\mathrm{dBm}/\mathrm{Hz}. $$

Then, the thermal noise power in the receiver input is:

$$ {P}_{in}=k{T}_0\cdotp N\cdotp {B}_n,\left[\mathrm{W}\right], $$

or in the logarithmic scale:

$$ {P}_{n. in}=k{T}_0+ NF+10\lg {B}_n=-174+5+10\lg \left(5\cdotp {10}^6\right)=-174+5+7+60=-102\;\mathrm{dBm}.\to \left(63.1\cdotp {10}^{-15}\left[\mathrm{W}\right]\right). $$

The equation for the noise factor through (5.11) can be written through the RF front-end gain in power and the average value of the own noise power:

$$ N=\frac{P_{s, in}/{P}_{n, in}}{K_{p, op.p}{P}_{s, in}/{K}_{p, op.p}\left({P}_{n, in}+{P}_{n, own}\right)}=1+\frac{P_{n, own}}{P_{n, in}} $$

(App. 5.4)

The value of the own noise power is calculated as:

P_{n, own} = P_{n, in}(N − 1).

For values NF = 5 dB (N = 3.16), the own noise power is:

Р_n,own = 63.1·10^-15(3.16-1) = 136.3·10^-15 [W]→ 0.136 [pW],

and the interception point coordinates are IP3 = 50 dBm = 10² [W].

The dynamic range on intermodulation distortions of the third order is calculated on the formula (5.75) correction completed:

$$ DB3=\frac{2\left( IP3-{P}_{n. own}\right)}{3}=\frac{2\left({10}^2-136\cdotp {10}^{-15}\right)}{3}=0.67\cdotp {10}^2=67\left[\mathrm{W}\right]\to 48.3\left[\mathrm{dBm}\right]. $$

The results obtained for the selected conditions show that to achieve the target value IIP3 = 50 dBm, it is sufficient that the dynamic range of receiver is 48.7 dBm, which is quaite simple to implement.